The invention is related to the field of combiners. More particularly, this invention relates to inline combiner networks which combine multiple frequency sources.
FIGS. 1 and 2 illustrate a combining network having two cavity resonators which uses intrusive coupling loops to couple signals from the different resonators. This approach has been used with ceramic, waveguide, and coaxial resonators. Coupling of a signal from each cavity is achieved in the following manner. A loop is placed into the cavity such that it couples into the magnetic field of the desired mode. The two loops (one for each cavity) are then joined at a common terminal and connected to the antenna port.
FIG. 3 shows a schematic of a general two-channel cavity combiner. The resonators are treated as a parallel LC resonator that is mutually coupled to two ports. The input port is connectedxe2x80x94usually through an isolatorxe2x80x94to a transmitter. The output port is connected to a junction via a transmission line, and a shunt component is attached at the junction to remove excess inductive reactance.
The resonator itself is used to pass the primary frequency while rejecting other frequencies by a certain amount.
The frequency response of a cavity centered at a frequency f0 is given in equation 1:                               H          ⁡                      (            f            )                          =                              (                          1              -                                                Q                  L                                                  Q                  U                                                      )                    ·                      1                                          1                +                                                      (                                          2                      ·                                              Q                        L                                            ·                                                                        f                          -                                                      f                            0                                                                                                    f                          0                                                                                      )                                    2                                                                                        (        1        )            
where QL is the ratio of the center frequency of the resonator to the frequency separation between the half-power (3 dB) points and is a function of the cavity coupling. The term QU is the unloaded Q of the resonator and represents the resonator Q if there was no external loading. The ratio of loaded Q to unloaded Q is the reflection coefficient at the center frequency of the resonator due to the internal losses of the resonator. The closer the ratio is to unity, the higher the loss in the cavity at midband. An important tradeoff in cavity performance is between narrow bandwidth and low loss.
The electrical length of the lines separating the resonators from the junction is determined from transmission-line theory. In transmission-line theory, it is widely known that an ideal line of length L transforms a load whose admittance is Y to an admittance YB such that:                               Y          B                =                              Y            0                    ·                                    (                                                                    cos                    ⁡                                          (                                              2                        ·                        π                        ·                                                  L                          λ                                                                    )                                                        ·                  Y                                +                                  1                  ⁢                                      i                    ·                                          sin                      ⁡                                              (                                                  2                          ·                          π                          ·                                                      L                            λ                                                                          )                                                              ·                                          Y                      0                                                                                  )                                      (                                                                    cos                    ⁡                                          (                                              2                        ·                        π                        ·                                                  L                          λ                                                                    )                                                        ·                                      Y                    0                                                  +                                  1                  ⁢                                      i                    ·                                          sin                      ⁡                                              (                                                  2                          ·                          π                          ·                                                      L                            λ                                                                          )                                                              ·                    Y                                                              )                                                          (        2        )            
where Y0 is the characteristic admittance of the transmission line, and xcex is the wavelength in the transmission line. This equation is found as equation 14 in Ramo, S; Whinnery, J.; Van Duzer, T.; Fields and Waves in Communications Electronics, 3rd Edition., 1994, John Wiley and Sons, New York, pp229-232, p254-256, hereby incorporated by reference. The transmission line can be several different shapes, such as coaxial or parallel wire. The embodiment we use uses a air-dielectric microstrip line designed such that the characteristic impedance Z0 is 50 ohms, which corresponds to a characteristic admittance Y0 of 1/Z0 or 0.02 mhos.
One of the well known property of ideal transmission lines is that the impedances tend to repeat themselves every half-wavelength. For example, a shorted transmission line (Yxe2x86x92∞) acts like an open circuit when the distance from the short is xcex/4xe2x80x94one quarter wavelength. When the distance reaches xcex/2xe2x80x94one half wavelengthxe2x80x94the admittance is that of short-circuit again. The impedance curves can be found in Pozar, D.; Microwave Engineering, 1993, Addison Wesley, New York, pp 76-84, hereby incorporated by reference. In the case where the admittance is Y, the transformed admittance YB is given in equation 3.                               Y          B                =                              Y            0            2                    Y                                    (        3        )            
Equation 3 shows that the quarter-wave transmission line acts as an admittance inverter because the higher admittances become low admittances at the opposite end of the transmission line.
The admittance of the isolated resonator loaded on the output with a load with admittance Y0 is approximately given as equation 4.                     Y        =                              Y            0                    ·                      (                          1              +                                                Q                  L                                                  Q                  U                                                      )                    ·                      (                          1              +                              2                ⁢                                  j                  ·                                      Q                    L                                    ·                                                            f                      -                                              f                        0                                                                                    f                      0                                                                                            )                                              (        4        )            
Equation 4 shows that the admittance Y becomes very large as the frequency f becomes more distant from f0. This means that an ideal parallel resonator becomes a short circuit at frequencies far from resonance, and a quarter-wave resonator will transform the near-short circuit.
Using the preferred embodiment as shown in FIG. 3, the resonators are set for center frequencies of f1 for the TX1 cavity and f2 for the TX2 cavity. In an ideal parallel-cavity resonator, the electrical length of the loop would be zero, and the cavity resonator""s off-resonance admittance would approach the infinite conductivity of a short circuit as the TX2 resonator frequency becomes further from f2. In such a case, attaching a transmission line of a quarter-wavelength would make the cavity look like a very low admittance and approach an open-circuit off the resonant frequency of the cavity at the other end of the cable.
If this admittance was placed in parallel with the antenna which is assumed to have an admittance of Y0, then the additional xe2x80x9cshuntingxe2x80x9d loss xcex1sh caused by the joined cavity is given in equation 5.                               α                      s            ⁢                          xe2x80x83                        ⁢            h                          =                  "LeftBracketingBar"                      2                          2              +                                                Y                  B                                                  Y                  0                                                              "RightBracketingBar"                                    (        5        )            
As the magnitude of the YB/Y0 ratio approaches zero, the shunting loss approaches zero. This is expected since an open circuit in parallel with any admittance has no effect on said admittance. If a second cavity on a frequency sufficiently separated from the first cavity is also attached to a quarter-wave transmission line, they can be joined to a common output. The first cavity on its resonant frequency only sees a small additional loading from the second cavity and vice versa.
As equation 4 shows, the cavity""s frequency response has an effect on the admittance off resonance or off the cavity""s resonant frequency. However, the combiner can still be used to combine cavities as long as the frequency separation between cavities is such that the response of one cavity frequency on the neighbor""s cavity response is down 4-6 dB from the center of the response. In such a case, the shunting loss approaches 1.3 dB. The shunting loss can be as high as 1.5 dB with multiple channels and still be useable in most systems where frequency separations are tight.
Ideally, the two loops in FIGS. 1 and 2 should be separated electrically from the junction by a transmission line whose length is one-quarter of a wavelength. In such a case, the shunt reactance shown in FIGS. 3 and 4 would be unnecessary. Unfortunately, an exact quarter-wave line is difficult to define or achieve. For example, all cavities have some small inductive reactance due to the finite length of the loop. FIG. 3 shows the general case where the line separating the cavities in the combiner is less thanxe2x80x94but fairly close toxe2x80x94a one-quarter-wavelength transmission line. The schematic includes the inductive reactance of the loop. Though not an exact quarter-wave line length, the two resonators can be connected as shown as long as the internal shunt reactance at the junction is cancelled using a shunt network. In the case where the separating lines are less than a quarter-wave in length, the internal shunt reactance at the junction is cancelled using a capacitor Cbal is shown in FIG. 3.
The main difficulty with using internal loops to couple signals from the cavity resonator is the electrical length required to reach the strong field regionxe2x80x94particularly in ceramic resonators. Because of the cavity size, the loop become so long that the lines are longer than quarter-wave. In the case where the lines are longer than a quarter-wavelength but less than a multiple of a half-wavelength, a shunt inductor is required to cancel the internal shunt reactance. In the case shown in FIG. 4, a fixed shunt inductor Lbal was chosen to be a fixed value and a shunt capacitor Cbal is placed across the inductor to electrically cancel the combined reactance of the balancing inductor and the residual reactance from the cavities and network. Further, the additional electrical length reduces the tuning range of the combiner because the lines are electrically longer and the inductorxe2x80x94usually implemented as a shorted transmission line stubxe2x80x94has a frequency dependence that further limits the useable range of the combiner.
Looking again at equation 1, YB equals Y whenever the cosine terms become 1 and the sine terms become zero. These occur at zero-length and at half-wavelength intervals. In the zero-length case, the two cavity outputs would be directly connected at the output, and the output signal from said cavity would be loaded down by the reactance and conductance of each adjacent cavity. A balancing capacitor can be addedxe2x80x94similar to what is shown in FIG. 3xe2x80x94but the cavities would still be, in essence, in parallel. As a result, more than half of the power going into one cavity would end up either reflected back or go directly into the adjacent cavity and out to the other input. This is a very undesirable condition. From equation one, it is seen that this condition also occurs if the cavities are combined using half-wavelength transmission lines. Again, there is no way to compensate this network. Consequently, it is preferable that the effective length from the cavity output to the junction not be a multiple of a half-wavelength. Thus, using a half-wave transmission line to couple energy from each cavity, the loops are effectively in parallel and there is low isolation between cavities.
Another issue with the loop design is that the only means of adjusting the coupling from the cavity is by adjusting the height of the loop. Sometimes, the loop has to be adjusted for optimal combiner/cavity performance. To make the adjustment, one has to loosen the ground side of the loop, move the ground up or down using a tool that protrudes into the cavity, retighten the locking hardware, and then make a measurement to determine if further adjustment is required. This approach is time consuming because the measurement is not accurate until the loop is tightened. In addition, sometimes the loop moves during the adjustment process. This results in the loop having to be adjusted additional times.
Another approach disclosed in the prior art was to use a common coaxial resonator to couple electromagnetic energy from each of the cavity resonators. A resulting standing wave in the common coaxial resonator couples into each cavity through apertures, one for each cavity resonator. The apertures are located a prescribed distance along the resonator transmission line as shown in a cut-away view in FIG. 5.
This approach works well if the electrical length between cavities is in half-wave increments. This is the case if the common resonator is a multiple half-wavelength coaxial resonator. In that case, the coaxial resonator""s length is a multiple half-wavelength of the average frequency of the combiner. Stated another way, the physical length of the coaxial resonator is a multiple half-wavelength of the average frequency of the input signal comprising a plurality of microwave signal frequencies output at the output port. Using half-wave increments, the signals are, effectively, combined in parallel. Therefore, the coaxial resonator appears as a low impedance to any of the input channel frequencies.
Unfortunately, in many cases there are restrictions on the length of the combiner such that that half-wave physical spacing is very difficult to achieve. Furthermore, the shunt reactance at the output junction or port would be difficult to predict. Consequently, a complicated compensating network would be needed to balance the phases of the different signals. In addition, low-loss combining would be difficult in that configuration.
Furthermore, even if there was enough room to electrically space the apertures by a half-wave, the outer channels would be very long electrically. For example, a six-channel unit would have its outer channels with 1.25 wavelengths between the aperture and the output. That would limit the bandwidth of the junction rather dramatically since only very high frequencies could be combined due to the reciprocal relationship between frequency and wavelength, i.e., the higher the frequency, the shorter the wavelength.
In a preferred embodiment, the invention is a combiner comprising a common port, a plurality of cavity resonators, a plurality of apertures and a combining mechanism operably connected to the common port and coupled to the plurality of resonators through apertures.
In another preferred embodiment, the combining mechanism comprises a junction to combine signals from a pair of cavity resonators. Transmission lines a quarter-wavelength or less in length connect the junction to the apertures.
In still another preferred embodiment, the invention comprises at least one edge pair of cavity resonators and a central pair of cavity resonators. The outputs of the edge pair of resonators are connected to a common port through half-wave transmission lines. The center pair of resonators are connected to the common port.
In still another preferred embodiment, the invention further comprises sliding covers located over the apertures to adjust coupling. A free-rotating screw adjusts the aperture by moving the sliding cover. The sliding covered is secured using at least one locking screw.